77GHz Down-Conversion Mixer with +18.4 dB High-Gain, +12.2 dBm OIP3, and Low-Noise in 90-nm CMOS Technology

77GHz Down-Conversion Mixer with +18.4 dB High-Gain, +12.2 dBm OIP3, and Low-Noise in 90-nm CMOS Technology | Research Square window.SnipcartSettings = { analytics: { enabled: false } }; (function() { var accessVector = localStorage.getItem('access_vector') || ''; window.dataLayer = window.dataLayer || []; if (accessVector) { window.dataLayer.push({ user: { profile: { profileInfo: { snid: accessVector } } } }); } })(); (function(w,d,s,l,i){w[l]=w[l]||[];w[l].push({'gtm.start':new Date().getTime(),event:'gtm.js'});var f=d.getElementsByTagName(s)[0],j=d.createElement(s),dl=l!='dataLayer'?'&l='+l:'';j.async=true;j.src='https://www.googletagmanager.com/gtm.js?id='+i+dl;f.parentNode.insertBefore(j,f);})(window,document,'script','dataLayer','GTM-K279D39R'); Browse Preprints In Review Journals COVID-19 Preprints AJE Video Bytes Research Tools Research Promotion AJE Professional Editing AJE Rubriq About Preprint Platform In Review Editorial Policies Our Team Advisory Board Help Center Sign In Submit a Preprint Cite Share Download PDF Research Article 77GHz Down-Conversion Mixer with +18.4 dB High-Gain, +12.2 dBm OIP3, and Low-Noise in 90-nm CMOS Technology Huan Zhang, Sida Tang, Mengye Cai, Yanfeng Jiang This is a preprint; it has not been peer reviewed by a journal. https://doi.org/ 10.21203/rs.3.rs-2373644/v1 This work is licensed under a CC BY 4.0 License Status: Published Journal Publication published 13 Apr, 2023 Read the published version in Journal of Infrared, Millimeter, and Terahertz Waves → Version 1 posted 7 You are reading this latest preprint version Abstract In radio-frequency (RF) transceiver system, the receiver is used to convert RF signal to low/medium frequency for signal processing and information gathering. In the paper, a novel high-gain, low-noise CMOS mixer is designed and analyzed. In the design, gain-boosting and PMOS dynamic switching current are employed to achieve better effects of noise cancellation and trans-conductance enhancement. In order to improve the mixer's performance, the designed structure is mainly based on the double-balanced Gilbert lattice mixer combined with the parallel LC resonant and optimum biasing networks. The mixer shows high conversion gain (CG), low noise figure (NF), and low power consumption performances based on made possible by 90-nm CMOS technology. Operated at 77 GHz, the input third-order intercept point (IIP3) is -6.14 dBm, and the maximum conversion gain is 18.4 dB. At an IF frequency of 200 MHz, a bilateral band noise figure of 9.2 dB is recorded, while 6.96 mW is consumed with 1.2 V and 2 dBm LO power. 77 GHz down-conversion mixer gain-boosting LC resonant network Figures Figure 1 Figure 2 Figure 3 Figure 4 Figure 5 Figure 6 Figure 7 Figure 8 Figure 9 Figure 10 Figure 11 Figure 12 1 Introduction The down-conversion mixer is a crucial component of the millimeter-wave receiver system. It can take the signal from the low noise amplifier (LNA) of the preamplifier module for amplification and demodulation. There are many key parameters to evaluate its performance, including power consumption, inter-port isolation, linearity, NF, CG, impedance matching, and the cost, etc. . In recent years, people have witnessed the development of high-frequency RF CMOS, and the next hot area of RF IC design will focus on V -band and beyond [ 1 ]. The potential applications include ultra-wideband long-range wireless networks, automotive image radar, and medical diagnosis instrumentation that all depend on high-speed information transmitting/receiving. Although CMOS passive mixers are becoming more and more common due to their high linearity and low flicker noise angle, the presence of conversion losses and inadequate reverse isolation in these mixers frequently results in dangerous NF and IQ crosstalk problems, necessitating additional complexity circuit modules to address the performance issues [ 2 ]. Contrarily, the Gilbert double-balanced mixer architecture is frequently utilized in millimeter-wave applications due to its compact layout, excellent port isolation properties, moderately high performance, and improved noise rejection for subsequent building blocks. However, the traditional Gilbert mixer shows the drawback of high NF and large power headroom, which materially reduces the total receiver performance. In particular, the local oscillator (LO) on/off switching phase of the mixer produces a considerable amount of flicker noise, which is conveyed to the output and reduces the transceiver's sensitivity. In the context of enhanced linearity and CG, the discharge structure of the circuit offers an interesting topology for a high-performance mixer [ 3 ]. In recent years, a number of studies have been reported on improving mixer noise. Major efforts have been made to suppress the indirect/direct noise mechanism [ 4 ] while enhancing the gain performance of the circuit by employing current bleeding and resonance techniques in the tail node of the switching pair. To achieve the low-noise, the mixer uses a noise-reduction trans-conductor with noise-canceling technology found in LNA [ 5 ]. In order to suppress the noise effect of LO switching pairs, the inductive source degradation technology is designed, which can obtain good noise performance by improving the linearity and input coupling [ 6 ]. The dynamic current bleeding technique is also proposed to improve the flicker-noise performance [ 7 ]. It injects a dynamic current during the LO switching event, which is equal to the bias current of the LO switches. Additionally, a capacitor cross-coupled (CCC) trans-conductance is used as the input stage to reduce the NF of the mixer while providing wideband input matching [ 8 ]. However, these structures still need to be improved for the balanced merits. In particulate, it is still challenging to balance noise and CG within low power restrictions while providing the growing bandwidth demands of multi-band and multimode transceivers. In the paper, a novel Gilbert down-conversion mixer is presented, which can work in 71 GHz to 86 GHz. In the design, multiple improvements, such as the good CG, the low noise, and low DC power consumption, are implemented by the adoption of a PMOS dynamic current switching method technique with a tuned inductor, an improved gain-boosting method, and an active load design. The tuned inductor L T and PMOS cross-coupled pairs serve as the dynamic current switching circuit topology. The branch currents can independently control the DC current during the switching stage and the trans-conductance stage. This method reduces the current passing through the switching stages and lessens the parasitic capacitance of the switching devices, by which improving the switching efficiency, and reducing the flicker noise while increasing the gain. To further improve the gain of the mixer, a modified gain-boosting circuit in the trans-conductance stage is used to increase CG and decrease NF, in which the g m and the output resistance of the amplifier are increased and the second-order harmonics are suppressed. A sine wave LO signal driver is used, which can be helpful for high gain and low noise. Additionally, the limitation of the switch-to-tail node parasitic capacitance C P of the mixer at high-frequency operation is substantially alleviated by the peak inductor L P . An IF buffer is added to the IF output stage to increase the gain, while can also separate the receive link from the digital baseband connection. The proposed mixer architecture is designed and simulated based on 90-nm CMOS standard technology. The paper is organized as follows. Section 2 describes the general circuit structure of the down-conversion mixer, with circuit design enhancements for the performance metrics of CG and NF. In Section 3, comparative simulation results for the mixer are provided. Section 4 presents the results. 2 Improvement Of The Cg And The Nf Of A Gilbert Mixer An active double-balanced Gilbert cell mixer is used for the investigation due to its high isolation between the LO and RF ports to eliminate the DC bias. Figure 1 shows the mixer's schematic diagram, including three primary components: the trans-conductance stage, the switching stage, and the load stage. The commutated RF current is down-converted to the IF voltage. To attain good CG and NF, the transistors should be operated in the saturation region. The overall circuit diagram of the 77-GHz CMOS Gilbert double-balanced downconverter mixer is shown in Fig. 2. It is composed of a trans-conductor pair of six transistors (M 1 -M 6 ) acting as the RF input stage, a differential LO input common-source stage transistor (M 7 -M 10 ) acting as the switching stage for switching the bias current, a pair of PMOS transistors (M 11 -M 12 ) constituting the active output load stage, and the IF buffer for the IF signal output. As shown in Fig. 2, current-controlled switching devices, and ideal transformers are used at each of the signal input ports to change the single-ended signals to the differential ones. The trans-conductance and the switching stages are matched to be 50 Ω impedance using an LC matching network with additional capacitors ( C pad ) and inductors ( L wire ). The structure is passive in order to decrease the active device loss and the NF. So CG is increased. Optimal biasing methods are employed at the input stage to improve the mixer's linearity while increasing the gain. The mixer's performance is sensitive to the LO power, particularly to the balance between the gain and the linearity. The two differential ports of T2 are linked in series with coupling capacitors C 3 , accounting for the inductive impedance of their inputs for better matching the inputs of the LO switching stages M 7 –M 10 . Additionally, C 3 is used to isolate the DC level between the center ground tap of T2 and the bias voltage of the switch to the transistor. In this way, the gate voltage swing of the switching device is immune to the LO power. The mixer's stability can be improved by preventing changes in 1-dB compression power (P 1dB ) and CG by the power. The designed parameters of the proposed mixer are listed in Table 1 . The mixer's conversion gain is a crucial parameter. The switching pair approximates an ideal switch when the LO signal is an ideal square wave. Ref. [ 9 ] provides the voltage conversion gain of a typical mixer when the LO signal is an ideal sine wave. $${A_{V,conv}} \approx \frac{2}{\pi }{g_m}{R_L}\left[ {1 - \frac{{\sqrt 2 {{\left( {VGS - VTH} \right)}_{sw}}}}{{\pi {V_{LO}}}}} \right]$$ 1 where (V GS -V TH ) sw the switching tube's overdrive voltage, R L the mixer load impedance, V LO the magnitude of the local oscillation signal, and g m the RF input stage's trans-conductance. In order to get the desired voltage gain of the mixer, increasing g m would be an effective way. However, high g m would increase the circuit's power consumption, and boost the load impedance R L , thus limiting the CG of the circuit. The C P charging and discharging currents of the common-source nodes P and Q appear as spike pulses when the LO signal amplitude is large. A strong LO signal causes the high leakage current. If the overdrive voltage of the switch device is reduced, the size of the switch device is increased to remain the same the current flowing through the device, resulting in a reduction in the mixer's performance. To keep the device be operated by LO signal, the transistor is biased near the edge of the NMOS triode area, i.e., V GS = V TH . The MOS device may reach a narrower gate width W because of the gain-boosting and noise-canceling techniques utilized in the trans-conductance stage construction. This can decrease parasitic capacitance and lower the overdrive voltage, which results in a reduction in voltage gain. Table 1 Component parameters of the proposed mixer. Device Parameter Device Parameter ( W/L ) 1,4 20µm /0.1µm R 1 /R 3 2 kΩ ( W/L ) 2,5 20µm /0.1µm R 2 100 Ω ( W/L ) 3,6 100µm /0.1µm R 4 40 kΩ ( W/L ) 7,10 46µm /0.1µm L P 100 pH ( W/L ) 11,12 60µm /0.2µm L T 70 pH ( W/L ) 13 120µm /0.2µm R L 500 Ω ( W/L ) 14,15 30µm /0.1µm V b1 0.7 V ( W/L ) 16,17 96µm /0.1µm V b2 0.85 V ( W/L ) 18,19 160µm /0.1µm V b3 1 V The suggested mixer, as shown in Fig. 3, employs an enhanced gain-boosting circuit based on the g m -boosting approach and the noise-canceling technique utilized in the LNA in Ref. [ 10 ]. Transistors M 1 -M 3 are used to convert the RF input voltage into RF current. During the first stage, the common-gate transistor is an input-matched network with an impedance of 1/g m . The input impedance of the trans-conductance stage is comparable to the input impedance of the common-gate amplifier when the noise caused by the common gate transistor is considered. A high input impedance and equivalent trans-conductance may be attained. $$Zin=\frac{{rO1}}{{1+\left( {gm1+gmb1} \right)rO1}}$$ 2 $$Gm \approx gm1gm2rO1+\frac{{gm3}}{{1+gm3R2+{{R2} \mathord{\left/ {\vphantom {{R2} {R3}}} \right. \kern-0pt} {R3}}}}$$ 3 M 3 is a common-source transistor with negative feedback at the source stage, allowing the output device to become more linear in order to make up for the mixer design's IIP3 performance. The current from the LO stage is utilized by M 3 as the trans-conductance current when the transistor's gate is connected with the RF input. Its gain is increased due to the high output impedance of the common-gate stage matching network. In addition, in order to increase the equivalent trans-conductance of the circuit and suppress the noise, a g m -boosting circuit is added after the matching circuit. The gm-boosting circuit consists of two amplifiers (M 2 , M 3 ) in a common-source configuration. The topology has a higher gain compared to the common-source amplifier by adding the input signal to the gate of transistor M 3 . $$Zout=\left[ {\left( {1+gm3rO3} \right)R2+rO3} \right]\parallel rO2$$ 4 $$AV,pro \approx \left[ {gm1gm2rO1+\frac{{gm3}}{{1+gm3R2+{{R2} \mathord{\left/ {\vphantom {{R2} {R3}}} \right. \kern-0pt} {R3}}}}} \right]\cdot \left[ {\left( {gm3rO3R2} \right)\parallel rO2} \right]$$ 5 To evaluate the sizes of M 2 and M 3 , high g m is necessary as it is proportional to the gain of the mixer, which can reduce the NF. So large M 3 is always required. However, big size could lead to large gate-source capacitance ( C GS ) and gate-drain capacitance ( C GD ), which could impact the circuit's input matching and increase the noise. Therefore, a PMOS dynamic current switching circuit with a tuned inductor is adopted in the design in the paper. When the LO signal is positive, the switch turns on transistors M 7 and M 9 , while turns off transistors M 8 and M 10 . During the second half cycle, the switches are opposite. During the switching, the varying v n modulates the switching time, and the v n moves directly to the IF output to affect the NF of the circuit. As seen in Fig. 4, noise can be generated when both switching devices are turned on simultaneously, or at the zero-crossing point. The switching noise is the main part of the mixer's whole noise performance. The output resistance can be increased by decreasing the bias current through the switching devices. This can lower the switching device's overdrive voltage without affecting the bias current through the trans-conductance stage. Dynamic Current-Bleeding (DCB) [ 11 ] is a method that can greatly improve noise performance as well as mixer gain performance. In the DCB approach, certain current is required to be injected at the moment of switching pair switching to eliminate the flicker noise at the output. Due to lower flicker noise angle of PMOS device, the PMOS based switching circuit can prevent the noise injection. It can sharpen the on/off of the LO switching stage by injecting current into the mixer core during the on/off switching. As shown in Fig. 5a, a similar fixed current injection is used in the traditional static current injection approach. It is ineffective for reducing the direct flicker noise since it creates more white noise and increases the effect of parasitic capacitance at the common-source node. If a dynamic current is injected at the common-source node, as shown in Fig. 5b, the problems encountered in injecting a fixed current can be avoided. However, there is still the influence of device capacitance on the noise of the circuit in Fig. 5b. To further enhance the NF of the PMOS switch current injection circuit, an efficient current injection technique is needed to lessen the injection of noise at all stages during the simultaneous conduction of the switching pair. In order to address these limitations and restrictions, a PMOS dynamic current injection approach with a tuned inductor L T is proposed, as shown in Fig. 5c. In Fig. 5c, a tuned inductor L T and a cross-coupled transistor pair (M 13 -M 15 ) made of PMOS transistors are constructed. The voltages at the P and Q nodes are varied with the input LO signal, which are controlled by M 14 and M 15 when the current I is injected into the common-source node. The branch current of the mixer switching pair is decreased by injecting current without altering the branch current of the trans-conductance stage during LO signal oscillation transition, so minimizing the noise pulses at the mixer's output IF port. To make sure that only the dynamic current can be injected with the LO signal at the zero-crossing point, a bias voltage design is needed. The bias voltage is small at the zero-crossing point. When the cross-coupled PMOS pair is injected, the top current source is turned on with current I flowing into the common-source node. Away from the zero-crossing point, due to the higher bias voltage, the PMOS device is disconnected. So there isn’t any injected current and the mixer works normally. With this method, the mixer's flicker noise is significantly reduced without adding more white noise. The mixer's linearity and bandwidth performance are not deteriorated at all. M 14 and M 15 need to be sized reasonably to inject enough current while avoiding extra parasitic capacitance to the switching pair's common-source end. In particular, the parasitic effect might produce a different flicker noise. Therefore, the tuned inductor L T is added in the PMOS cross-coupling pair to balance the PMOS device's device capacitance and enhance the mixer circuit's NF. According to the final testing findings, the circuit with the inductor shows 3 dB better NF than the other ones. Figure 6 shows the effects of the parasitic capacitors on the device. C GS in the saturation zone can influence the frequency response. The inter-stage parasitic capacitance C P can bypass some signal current to ground, which is another factor contributing to the considerable deterioration of the common-source common-gate amplifier in the millimeter-wave region. The high-frequency operating characteristics of the mixer are substantially controlled by the parasitic capacitance at the tail nodes P and Q of the switch pair, as illustrated in Fig. 2, which also has a considerable influence on the mixer's NF and gain. Figure 7a shows typical method for removing parasitic capacitance by adding an inter-stage series inductor L S . A π-shaped matching network is created by L S with the parasitic capacitors C P1 and C P2 . In fact, since the difference between the common-source output resistance r O3 and the common-gate input resistance 1/g m is so considerable, it is difficult to match with only one inductor and two parasitic capacitors. Employing two series inductors for differential circuit design would need extra chip area. A series-parallel inter-stage hybrid LC network is shown in Fig. 7b. The signal current shunted by C P2 is reduced by resonating L P with C P2 . The source of M 7 has a tail node impedance Z P of: $$Zp=rO3\parallel sLP\parallel \frac{1}{{j\omega LOCP}}$$ 6 where r O3 represents the small signal output resistance of transistor M 3 . Since the LO signal drives the switching device set in the saturation operation region, its Z P is quite small. Practically, resonance with C P2 requires a quite big inductor with large area. In addition, the PMOS’ nonlinear capacitance tends to produce harmonics and create leakage for the insertion current with the increment of the PMOS size. An improved LC resonant network with the resonant point at the LO frequency of 77.2 GHz is designed, as shown in Fig. 7c. A peak inductor L P at the common-source end of the LO switching devices is added to improve the noise performance. It can reduce the flicker noise by canceling the influence of C P2 between the switching and trans-conductance stages, which also increases the CG due to the lack of signal leakage through the parasitic capacitance. The parallel inductor L T is the tuned inductor in the dynamic current injection module. The LC parallel resonant network with the inter-stage parasitic capacitor C P1 can neutralize the parasitic capacitance of the PMOS cross-coupling to the circuit. The tuned circuit can suppress the harmonics and the leakage current caused by the inductive filling effect in the nonlinear capacitance. The RF high-impedance circuit by the inductor and C P at circuit nodes P and Q can decrease the loss of the switching circuit near the RF frequency. The indirect formation of flicker noise by the LO big signal charging and discharging low resistance channel can be reduced due to the closer frequency of the RF and LO signals. The proposed mixer circuit in the paper is assisted by the L P and L T resonance effects, with Z P being approximated by r O3 . The higher impedance of the switching stage corresponds to better common mode interference rejection at the RF input port. The inductor L P needs to be optimized for the design, and its value is chosen by maximum linearity and minimum NF. The peak gain of the mixer increases by roughly 12 dB and the noise decreases by around 9 dB when the inter-stage parallel inductors L T and L P are added. PMOS cross pairs are utilized as active loads to reduce the load-level noise. The system noise is increased by the current injection module of the circuit. The cross-coupling approach is used to mitigate the noise deterioration. The gate length L of the MOS device can be chosen as large as necessary to reduce the flicker noise of the transistor and to lessen its channel modulation impact, in favor of increasing the output impedance and the gain of the mixer. As long as the buffer circuit itself does not introduce excessive signal attenuation, no additional noise can be generated due to the high gain of the mixer before the IF buffer circuit. A self-biasing inverting amplifier is used as the IF output buffer circuit, including M 10 -M 13 and R 4 . The load circuit’s output signal swing is considerable, and the buffer needs a sizable enough linear range to guard the mixer’s linearity. The input and output DC levels can be expressed as: $$Vin=Vout=\frac{{(VDD+VTHP+VTHN)}}{2}$$ 7 The transistor's stability factor is defined to be K > 1 over the whole frequency range. It is feasible to improve the output gain by adjusting the value of R 4 . To keep the amplifier's front stage from being impacted by the input signal's DC level, the off-chip blocking capacitors C 4 is used. The buffer circuit can reduce the complexity of the circuit since no additional bias circuit is needed. The flicker noise and the channel modulation effect can be minimized by increasing the gate length L and gate width W of MOS devices. Filtering is necessary to obtain the IF signal because fundamental signals and harmonic signals can leak from the RF and LO ports to the IF side. Capacitor filtering is employed since each harmonic signal's frequency is far from that of the zero IF signal. 3 Simulation Results The mixer proposed in this research is implemented in TSMC 90-nm RF CMOS process and simulated with ADS (Advanced Design System). Figure 8a shows the way the overall circuit of the mixer is implemented. The RF and LO ports convert single-ended to differential signals using an external ideal transformer during simulation. Both the trans-conductance stage of the mixer and the LO switching stage employ the same load resistance and same bias for a fair comparison. Figure 8b shows a micrograph of the circuit with a chip area of 0.5 × 0.5mm 2 for the core circuit of the mixer. The simulation results for the mixer with various inter-stage network architectures are compiled in Fig. 9a and Fig. 9b, respectively. Figure 9a shows the simulation results of the LO input power versus the conversion gain. With RF frequencies 71–86 GHz and IF frequencies at 200 MHz, the mixer's CG varies from 12.6 dB to 18.4 dB. The maximum CG is achieved when LO power is 2 dBm. The PMOS dynamic current injection can increase the mixer's voltage gain. The mixer shows CG of 18.4 dB due to the addition of the inductor L P and the improved gain-boosting circuit in the trans-conductance stage. As shown in Fig. 9b, the addition of the inductor L P lowers the mixer's SSB NF by 2.6 dB. The SSB NF is decreased by around 3.3 dB by adopting a dynamic current injection circuit with tuned inductor L T , with a final SSB NF of 12.2 dB. As shown in Fig. 10a, with the frequency from 71 GHz to 86 GHz, the reflection coefficient S 11 is less than − 40 dB at the RF port, and the transmission coefficient S 21 is less than − 30 dB at the IF port. Figure 10b displays the test results for the LO-to-IF port, LO-to-RF port, and RF-to-IF port isolation, with isolation values larger than 50 dB, 60 dB, and 45 dB, respectively. The mixer also provides enough linearity to maintain high-gain and low-noise. The IP 1dB and the IIP3 at RF frequency of 77 GHz are − 17.2 and − 6.14 dBm, respectively. Figure 11a shows simulated and measured conversion gain and NF versus LO power, with fixed LO of 77.2 GHz and RF of 77 GHz. Figure 11b shows the relationship between IF output power and RF input power. Figure 12a shows the relationship between RF input power and the conversion gain. Figure 12b shows the impact of various IF frequencies on the gain performance with 40 dBm RF power. The mixer’s core circuit consumes around 6 mA of current at a supply voltage of 1.2 V. Fig. 12 (a) Simulation results of the RF input power versus the conversion gain. (b) Simulation results of conversion gain at different IF frequencies. A figure-of-merit (FOM) suitable for evaluating the performance of a down-conversion mixer, a low-noise amplifier, or a receiver front-end can be defined as below [ 12 ]: $$FOM1=10\log \left( {\frac{{{{10}^{CG[dB]/20}} \cdot {{10}^{\left( {IIP3[dBm] - 10} \right)/20}}}}{{{{10}^{NF[dB]/10}} \cdot PDC[mW]}} \cdot \frac{{fRF}}{{1GHz}}} \right)$$ 8 where CG [dB] the average CG in magnitude; IIP3 [dBm] represents the input third-order intercept point, NF [dB] the DSB NF in magnitude, P DC [mW] power dissipation in milliwatts, f RF the value of the RF input frequency. The FOM shown in Eq. 8 includes the most relevant parameters for evaluating down-conversion mixer for low-power, high-gain, low-noise, and high-linearity applications. Moreover, in numerous pieces of previous work, the information of NF and linearity are not available. To compare with them, the following simplified FOM can be used. $$FOM2=10\log \left( {\frac{{{{10}^{CG[dB]/20}}}}{{PDC[mW]}} \cdot \frac{{fRF}}{{1GHz}}} \right)$$ 9 Table 2 summarizes the performance of the proposed down-conversion mixer and compares the parameters with the recently reported mixers. Due to the application of an improved gain-boosting approach, the proposed mixer exhibits a significant enhancement in CG and the noise compared with the mixers published in [ 13 ], [ 14 ]. Comparing with the 76–77 GHz CMOS mixer in [ 9 ], the proposed mixer shows better CG, larger port-to-port isolation, and larger FOM value. The proposed mixer consumes less power than the mixers described in [ 9 ], [ 14 ], and [ 15 ]. In comparison to the mixer described in [ 16 ], the proposed mixer offers better noise performance. It can be concluded that the proposed mixer exhibits a competitive performance with the low-power, the high-gain and the low-noise performance. 4 Conclusions In the paper, a novel mixer is designed, with the improved gain and better noise performance. In the design, an improved trans-conductance circuit and a PMOS dynamic current switching technique with inductance are adopted. At the input transistor's trans-conductance stage, the gain-boosting approach is used to increase the trans-conductance and the output resistance. At the same time, the circuit noise can be suppressed effectively. The second-order derivative trans-conductance is kept zero in the input transistor by using the optimal biasing technique, which enhances the linearity without decreasing the gain. To further increase the mixer’s gain and better NF, a dynamic current injection circuit with a tuned inductor is added at the LO input stage. Simulation results based on TSMC 90-nm CMOS process show that the proposed mixer consumes only 6.96 mW with 1.2 V supply, and DSB NF of 9.2 dB, CG of + 18.4 dB, and OIP3 of 12.2 dBm. The proposed mixer shows potential application in the field of vehicle radar system, providing the benefits of high-gain, low-noise, and low power consumption, etc. . Table 2 Summary results and performance comparison of other mixers. Performance parameter References This work [ 9 ] [ 13 ] [ 14 ] [ 15 ] [ 16 ] Technology 90-nm CMOS 65-nm CMOS 90-nm CMOS 90-nm CMOS 0.18-µm CMOS 90-nm CMOS Freq. (GHz) 71–86 75–90 70–88 75–85 62–85 72–100 CG (dB) 18.4 5 5.3-9 1.5 -4.3 17.1 NF (dB) 9.2 N/A 22 23.3 N/A 16.4 IP 1dB (dBm) -17.2 2.5 -13 -9 2.14 -13 OIP3 (dBm) 12.23 N/A N/A N/A N/A 14.1 P LO (dBm) 2 0 -4 5 5 0 ISO. (dB) > 45 37 > 40 a 49.2 30 43.3 Size (mm 2 ) 0.25 0.14 0.32 0.62 0.38 0.61 V DD (V) 1.2 1.5 1.2 1.2 1.2 1.2 P DC (mW) 6.96 b /10.56 8 b /12 5 13 10.8 8.6 FOM 1 2.36 N/A -11.94 -18.3 N/A -4.98 FOM 2 19.64 12.28 15.5 9.4 c 8.6 18.1 a 2LO- to-RF Isolation b Without buffer c At 80 GHz Declarations Ethics Approval and Consent to Participate Not applicable. Competing interests The authors declare no competing interests. Authors' contributions Huan Zhang and Yanfeng Jiang wrote the main manuscript text; Sida Tang and Mengye Cai assisted with measurements. All authors reviewed the manuscript and contributed with comments. Funding Open access funding provided by The Natural Science Foundation of China with No.61774078. Availability of data and materials The data that support the findings of this study are available from the corresponding author upon reasonable request. 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Wu Y-C, Wang H: An E-band Double-Balanced Subharmonic Mixer With High Conversion Gain and Low Power in 90-nm CMOS Process. IEEE Microwave and Wireless Components Letters 2018, 28 : 70-72, https:// doi. org/ 10.1109/ LMWC.2017.2779847. Lin YS, Li GH: 13 mW 80 GHz down‐conversion mixer with 1.5 dB gain and 49.2 dB LO–RF isolation. Electronics Letters 2014, 50 : 1449-1451, https:// doi. org/ 10.1049/ el.2014.1263. Chen Z, Liu H, Liu Z, Jiang Z, Yu Y, Wu Y, Zhao C, Kang K: A 62–85-GHz High Linearity Upconversion Mixer With 18-GHz IF Bandwidth. IEEE Microwave and Wireless Components Letters 2019, 29 : 219-221, https:// doi. org/ 10.1109/ LMWC.2019.2894979. Lin Y-S, Lan K-S: A W-band CMOS down-conversion mixer using CMOS-inverter-based RF GM stage for conversion gain and linearity enhancement. Analog Integrated Circuits and Signal Processing 2019, 99 : 133-146, https:// doi. org/ 10.1109/ RWS.2019.8714325. Additional Declarations No competing interests reported. Cite Share Download PDF Status: Published Journal Publication published 13 Apr, 2023 Read the published version in Journal of Infrared, Millimeter, and Terahertz Waves → Version 1 posted Editorial decision: Major revision 04 Mar, 2023 Reviews received at journal 15 Jan, 2023 Reviewers agreed at journal 27 Dec, 2022 Reviewers invited by journal 27 Dec, 2022 Editor assigned by journal 26 Dec, 2022 Submission checks completed at journal 15 Dec, 2022 First submitted to journal 13 Dec, 2022 You are reading this latest preprint version Research Square lets you share your work early, gain feedback from the community, and start making changes to your manuscript prior to peer review in a journal. As a division of Research Square Company, we’re committed to making research communication faster, fairer, and more useful. We do this by developing innovative software and high quality services for the global research community. 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Also discoverable on Platform About Our Team In Review Editorial Policies Advisory Board Help Center Resources Author Services Accessibility API Access RSS feed Manage Cookie Preferences © Research Square 2026 | ISSN 2693-5015 (online) Privacy Policy Terms of Service Do Not Sell My Personal Information {"props":{"pageProps":{"initialData":{"identity":"rs-2373644","acceptedTermsAndConditions":true,"allowDirectSubmit":false,"archivedVersions":[],"articleType":"Research Article","associatedPublications":[],"authors":[{"id":160298244,"identity":"aad897b1-bad1-4b76-9f42-164ff3358cf0","order_by":0,"name":"Huan Zhang","email":"","orcid":"","institution":"Jiangnan University","correspondingAuthor":false,"submittingAuthor":false,"prefix":"","firstName":"Huan","middleName":"","lastName":"Zhang","suffix":""},{"id":160298246,"identity":"aac730d3-ce7e-4e44-add3-5970313f1b9f","order_by":1,"name":"Sida Tang","email":"","orcid":"","institution":"Wuxi University","correspondingAuthor":false,"submittingAuthor":false,"prefix":"","firstName":"Sida","middleName":"","lastName":"Tang","suffix":""},{"id":160298248,"identity":"bb602429-e9b9-4031-aed0-48ce39751a2b","order_by":2,"name":"Mengye Cai","email":"","orcid":"","institution":"Jiangnan University","correspondingAuthor":false,"submittingAuthor":false,"prefix":"","firstName":"Mengye","middleName":"","lastName":"Cai","suffix":""},{"id":160298249,"identity":"54fa2eca-ed63-4345-9933-db76febdca89","order_by":3,"name":"Yanfeng Jiang","email":"data:image/png;base64,iVBORw0KGgoAAAANSUhEUgAAAZAAAAAyAQMAAABI0h/eAAAABlBMVEX///8AAABVwtN+AAAACXBIWXMAAA7EAAAOxAGVKw4bAAAA4klEQVRIiWNgGAWjYBACxmYwJScHZYCBATFajI2J1wIFxokNSDz8WpjbmZ89/NpmkN4MZuywy2Ngb94mwVBzB4/D2MyNZdsMchvBjDPJxQw8x8okGI49w+cXM2nJtj9ALWAGc2KDRI6ZBGPDYTxa2L8BVRqkQxn1iQ3ybwhp4TGT/NhmkABlHAbawkNQS5k0wzkDw0YQg7HteGIbT1qxRcIx3FoM+49vk/xRZiAPZvxsq07sZz+88caHGjxaGoABzcsGZfAARdhAwgk4NTAwyIMc9+MPjIFH5SgYBaNgFIxcAAAZH06SHfiGhwAAAABJRU5ErkJggg==","orcid":"","institution":"Jiangnan University","correspondingAuthor":true,"submittingAuthor":false,"prefix":"","firstName":"Yanfeng","middleName":"","lastName":"Jiang","suffix":""}],"badges":[],"createdAt":"2022-12-13 11:14:22","currentVersionCode":1,"declarations":"","doi":"10.21203/rs.3.rs-2373644/v1","doiUrl":"https://doi.org/10.21203/rs.3.rs-2373644/v1","draftVersion":[],"editorialEvents":[{"content":"https://doi.org/10.1007/s10762-023-00917-2","type":"published","date":"2023-04-13T20:28:51+00:00"}],"editorialNote":"","failedWorkflow":false,"files":[{"id":30522535,"identity":"c933f4c0-b2f8-46b4-a05f-831ff2196201","added_by":"auto","created_at":"2022-12-19 15:34:18","extension":"png","order_by":1,"title":"Figure 1","display":"","copyAsset":false,"role":"figure","size":54432,"visible":true,"origin":"","legend":"\u003cp\u003eThe proposed improved active mixer architecture (the bias circuit not shown)\u003c/p\u003e","description":"","filename":"floatimage1.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/8b16b9bf854df57f30d002c4.png"},{"id":30525283,"identity":"b91bb3ad-8315-4a50-aaf1-99347ffb9d1b","added_by":"auto","created_at":"2022-12-19 16:06:19","extension":"png","order_by":2,"title":"Figure 2","display":"","copyAsset":false,"role":"figure","size":152836,"visible":true,"origin":"","legend":"\u003cp\u003eBlock diagram of double-balanced active mixer with bias circuit and IF Buffer stage\u003c/p\u003e","description":"","filename":"floatimage2.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/106e9077651b410ea30bff59.png"},{"id":30523347,"identity":"23b43cdb-dbdb-44eb-b3c3-9e86fb6ea970","added_by":"auto","created_at":"2022-12-19 15:42:19","extension":"png","order_by":3,"title":"Figure 3","display":"","copyAsset":false,"role":"figure","size":29125,"visible":true,"origin":"","legend":"\u003cp\u003eHalf circuit of the proposed improved differential trans-conductance stage.\u003c/p\u003e","description":"","filename":"floatimage3.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/f25d0b2756f6853af19c1db4.png"},{"id":30522537,"identity":"e99127a3-81b6-4df9-9a74-28df258420d8","added_by":"auto","created_at":"2022-12-19 15:34:19","extension":"png","order_by":4,"title":"Figure 4","display":"","copyAsset":false,"role":"figure","size":90785,"visible":true,"origin":"","legend":"\u003cp\u003eEffect of non-ideal signal LO signal on a mixer.\u003c/p\u003e","description":"","filename":"floatimage4.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/eff9a86420f30465ec8bcaf0.png"},{"id":30523348,"identity":"e59529e7-67ff-46de-9a73-ac50e25a392c","added_by":"auto","created_at":"2022-12-19 15:42:19","extension":"png","order_by":5,"title":"Figure 5","display":"","copyAsset":false,"role":"figure","size":57740,"visible":true,"origin":"","legend":"\u003cp\u003eCurrent injection technique (a) Traditional static current injection structure. (b) Effect of the parasitic capacitance in dynamic current injection circuits. (c) Proposed PMOS dynamic current injection circuit with a tuned inductor.\u003c/p\u003e","description":"","filename":"floatimage5.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/e36a23c9c6ef9d23842335b2.png"},{"id":30524986,"identity":"7618349f-5a8e-4119-ba66-f70718d3a6b9","added_by":"auto","created_at":"2022-12-19 15:58:19","extension":"png","order_by":6,"title":"Figure 6","display":"","copyAsset":false,"role":"figure","size":29478,"visible":true,"origin":"","legend":"\u003cp\u003eThe effect of parasitic capacitance on devices\u003c/p\u003e","description":"","filename":"floatimage6.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/b21dc729e642ce8a7a607b24.png"},{"id":30524043,"identity":"4c74d1ca-338c-4521-a595-b1d948f79d48","added_by":"auto","created_at":"2022-12-19 15:50:19","extension":"png","order_by":7,"title":"Figure 7","display":"","copyAsset":false,"role":"figure","size":69348,"visible":true,"origin":"","legend":"\u003cp\u003eInductance techniques for (a) Series inter-stage inductors. (b) Series-parallel hybrid inductors. (c) The proposed modified LC resonant network\u003c/p\u003e","description":"","filename":"floatimage7.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/8740ba2e95707c7703275f83.png"},{"id":30522543,"identity":"e5b8a430-508c-419f-ac34-e845c33608df","added_by":"auto","created_at":"2022-12-19 15:34:19","extension":"png","order_by":8,"title":"Figure 8","display":"","copyAsset":false,"role":"figure","size":215361,"visible":true,"origin":"","legend":"\u003cp\u003e(a) The way the overall circuit of the mixer is implemented. (b) Chip photograph of the mixer with chip size of 0.5 × 0.5 mm\u003csup\u003e2\u003c/sup\u003e.\u003c/p\u003e","description":"","filename":"floatimage8.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/435ad015571b8ca712d7634a.png"},{"id":30522545,"identity":"d1ac872c-0b4a-466d-ac7c-e342fc306bf0","added_by":"auto","created_at":"2022-12-19 15:34:19","extension":"png","order_by":9,"title":"Figure 9","display":"","copyAsset":false,"role":"figure","size":141064,"visible":true,"origin":"","legend":"\u003cp\u003e(a) Simulation results of the LO input power versus the conversion gain. (b) Simulation results of the LO input power versus SSB NF\u003c/p\u003e","description":"","filename":"floatimage9.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/6d5e70e89b35c35d2138ceb1.png"},{"id":30523354,"identity":"31cfae62-1d6e-43d4-aac9-6ec23e238174","added_by":"auto","created_at":"2022-12-19 15:42:19","extension":"png","order_by":10,"title":"Figure 10","display":"","copyAsset":false,"role":"figure","size":92228,"visible":true,"origin":"","legend":"\u003cp\u003e(a) Input return losses S\u003csub\u003e11\u003c/sub\u003e and insertion losses S\u003csub\u003e21\u003c/sub\u003e versus the mixer input frequency. (b) Simulation results of the isolation degree of each mixer port.\u003c/p\u003e","description":"","filename":"floatimage10.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/271d426d1cdaa17721e9ab03.png"},{"id":30525282,"identity":"f676671e-7496-45e2-a268-36b4fe58e1cb","added_by":"auto","created_at":"2022-12-19 16:06:19","extension":"png","order_by":11,"title":"Figure 11","display":"","copyAsset":false,"role":"figure","size":102285,"visible":true,"origin":"","legend":"\u003cp\u003eFig. 1 \u0026nbsp;(a) The simulated and measured conversion gain and NF versus LO power. (b) IIP3 (\u003cem\u003ef\u003c/em\u003e\u003csub\u003e\u003cem\u003eLO\u003c/em\u003e\u003c/sub\u003e = 77.2 GHz, \u003cem\u003ef\u003c/em\u003e\u003csub\u003e\u003cem\u003eRF \u003c/em\u003e\u003c/sub\u003e= 77 GHz) and P1dB measurement results.\u003c/p\u003e","description":"","filename":"floatimage11.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/348d87efec5efaa45ef85980.png"},{"id":30524045,"identity":"55a8d798-34f8-4881-8c53-64d58e3407e3","added_by":"auto","created_at":"2022-12-19 15:50:19","extension":"png","order_by":12,"title":"Figure 12","display":"","copyAsset":false,"role":"figure","size":79488,"visible":true,"origin":"","legend":"\u003cp\u003e(a) Simulation results of the RF input power versus the conversion gain. (b) Simulation results of conversion gain at different IF frequencies\u003c/p\u003e","description":"","filename":"floatimage12.png","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/2854b6a5381c5429b77ca1b0.png"},{"id":44726819,"identity":"e0361181-5d2c-48a3-825b-be1a40f65a6c","added_by":"auto","created_at":"2023-10-16 20:49:05","extension":"pdf","order_by":0,"title":"","display":"","copyAsset":false,"role":"manuscript-pdf","size":1364701,"visible":true,"origin":"","legend":"","description":"","filename":"manuscript.pdf","url":"https://assets-eu.researchsquare.com/files/rs-2373644/v1/c360426f-987a-42af-8393-b4bffc19880d.pdf"}],"financialInterests":"No competing interests reported.","formattedTitle":"77GHz Down-Conversion Mixer with +18.4 dB High-Gain, +12.2 dBm OIP3, and Low-Noise in 90-nm CMOS Technology","fulltext":[{"header":"1 Introduction","content":"\u003cp\u003eThe down-conversion mixer is a crucial component of the millimeter-wave receiver system. It can take the signal from the low noise amplifier (LNA) of the preamplifier module for amplification and demodulation. There are many key parameters to evaluate its performance, including power consumption, inter-port isolation, linearity, NF, CG, impedance matching, and the cost, \u003cem\u003eetc.\u003c/em\u003e. In recent years, people have witnessed the development of high-frequency RF CMOS, and the next hot area of RF IC design will focus on \u003cem\u003eV\u003c/em\u003e-band and beyond [\u003cspan citationid=\"CR1\" class=\"CitationRef\"\u003e1\u003c/span\u003e]. The potential applications include ultra-wideband long-range wireless networks, automotive image radar, and medical diagnosis instrumentation that all depend on high-speed information transmitting/receiving.\u003c/p\u003e \u003cp\u003eAlthough CMOS passive mixers are becoming more and more common due to their high linearity and low flicker noise angle, the presence of conversion losses and inadequate reverse isolation in these mixers frequently results in dangerous NF and IQ crosstalk problems, necessitating additional complexity circuit modules to address the performance issues [\u003cspan citationid=\"CR2\" class=\"CitationRef\"\u003e2\u003c/span\u003e]. Contrarily, the Gilbert double-balanced mixer architecture is frequently utilized in millimeter-wave applications due to its compact layout, excellent port isolation properties, moderately high performance, and improved noise rejection for subsequent building blocks. However, the traditional Gilbert mixer shows the drawback of high NF and large power headroom, which materially reduces the total receiver performance. In particular, the local oscillator (LO) on/off switching phase of the mixer produces a considerable amount of flicker noise, which is conveyed to the output and reduces the transceiver's sensitivity. In the context of enhanced linearity and CG, the discharge structure of the circuit offers an interesting topology for a high-performance mixer [\u003cspan citationid=\"CR3\" class=\"CitationRef\"\u003e3\u003c/span\u003e]. In recent years, a number of studies have been reported on improving mixer noise. Major efforts have been made to suppress the indirect/direct noise mechanism [\u003cspan citationid=\"CR4\" class=\"CitationRef\"\u003e4\u003c/span\u003e] while enhancing the gain performance of the circuit by employing current bleeding and resonance techniques in the tail node of the switching pair. To achieve the low-noise, the mixer uses a noise-reduction trans-conductor with noise-canceling technology found in LNA [\u003cspan citationid=\"CR5\" class=\"CitationRef\"\u003e5\u003c/span\u003e]. In order to suppress the noise effect of LO switching pairs, the inductive source degradation technology is designed, which can obtain good noise performance by improving the linearity and input coupling [\u003cspan citationid=\"CR6\" class=\"CitationRef\"\u003e6\u003c/span\u003e]. The dynamic current bleeding technique is also proposed to improve the flicker-noise performance [\u003cspan citationid=\"CR7\" class=\"CitationRef\"\u003e7\u003c/span\u003e]. It injects a dynamic current during the LO switching event, which is equal to the bias current of the LO switches. Additionally, a capacitor cross-coupled (CCC) trans-conductance is used as the input stage to reduce the NF of the mixer while providing wideband input matching [\u003cspan citationid=\"CR8\" class=\"CitationRef\"\u003e8\u003c/span\u003e]. However, these structures still need to be improved for the balanced merits. In particulate, it is still challenging to balance noise and CG within low power restrictions while providing the growing bandwidth demands of multi-band and multimode transceivers.\u003c/p\u003e \u003cp\u003eIn the paper, a novel Gilbert down-conversion mixer is presented, which can work in 71 GHz to 86 GHz. In the design, multiple improvements, such as the good CG, the low noise, and low DC power consumption, are implemented by the adoption of a PMOS dynamic current switching method technique with a tuned inductor, an improved gain-boosting method, and an active load design. The tuned inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e and PMOS cross-coupled pairs serve as the dynamic current switching circuit topology. The branch currents can independently control the DC current during the switching stage and the trans-conductance stage. This method reduces the current passing through the switching stages and lessens the parasitic capacitance of the switching devices, by which improving the switching efficiency, and reducing the flicker noise while increasing the gain. To further improve the gain of the mixer, a modified gain-boosting circuit in the trans-conductance stage is used to increase CG and decrease NF, in which the g\u003csub\u003em\u003c/sub\u003e and the output resistance of the amplifier are increased and the second-order harmonics are suppressed.\u003c/p\u003e \u003cp\u003eA sine wave LO signal driver is used, which can be helpful for high gain and low noise. Additionally, the limitation of the switch-to-tail node parasitic capacitance \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP\u003c/sub\u003e of the mixer at high-frequency operation is substantially alleviated by the peak inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e. An IF buffer is added to the IF output stage to increase the gain, while can also separate the receive link from the digital baseband connection.\u003c/p\u003e \u003cp\u003eThe proposed mixer architecture is designed and simulated based on 90-nm CMOS standard technology. The paper is organized as follows. Section 2 describes the general circuit structure of the down-conversion mixer, with circuit design enhancements for the performance metrics of CG and NF. In Section 3, comparative simulation results for the mixer are provided. Section 4 presents the results.\u003c/p\u003e"},{"header":"2 Improvement Of The Cg And The Nf Of A Gilbert Mixer","content":"\u003cp\u003eAn active double-balanced Gilbert cell mixer is used for the investigation due to its high isolation between the LO and RF ports to eliminate the DC bias. Figure\u0026nbsp;1 shows the mixer's schematic diagram, including three primary components: the trans-conductance stage, the switching stage, and the load stage. The commutated RF current is down-converted to the IF voltage. To attain good CG and NF, the transistors should be operated in the saturation region.\u003c/p\u003e \u003cp\u003eThe overall circuit diagram of the 77-GHz CMOS Gilbert double-balanced downconverter mixer is shown in Fig.\u0026nbsp;2. It is composed of a trans-conductor pair of six transistors (M\u003csub\u003e1\u003c/sub\u003e-M\u003csub\u003e6\u003c/sub\u003e) acting as the RF input stage, a differential LO input common-source stage transistor (M\u003csub\u003e7\u003c/sub\u003e-M\u003csub\u003e10\u003c/sub\u003e) acting as the switching stage for switching the bias current, a pair of PMOS transistors (M\u003csub\u003e11\u003c/sub\u003e-M\u003csub\u003e12\u003c/sub\u003e) constituting the active output load stage, and the IF buffer for the IF signal output. As shown in Fig.\u0026nbsp;2, current-controlled switching devices, and ideal transformers are used at each of the signal input ports to change the single-ended signals to the differential ones. The trans-conductance and the switching stages are matched to be 50 Ω impedance using an LC matching network with additional capacitors (\u003cem\u003eC\u003c/em\u003e\u003csub\u003epad\u003c/sub\u003e) and inductors (\u003cem\u003eL\u003c/em\u003e\u003csub\u003ewire\u003c/sub\u003e). The structure is passive in order to decrease the active device loss and the NF. So CG is increased. Optimal biasing methods are employed at the input stage to improve the mixer's linearity while increasing the gain.\u003c/p\u003e \u003cp\u003eThe mixer's performance is sensitive to the LO power, particularly to the balance between the gain and the linearity. The two differential ports of T2 are linked in series with coupling capacitors \u003cem\u003eC\u003c/em\u003e\u003csub\u003e3\u003c/sub\u003e, accounting for the inductive impedance of their inputs for better matching the inputs of the LO switching stages M\u003csub\u003e7\u003c/sub\u003e\u0026ndash;M\u003csub\u003e10\u003c/sub\u003e. Additionally, \u003cem\u003eC\u003c/em\u003e\u003csub\u003e3\u003c/sub\u003e is used to isolate the DC level between the center ground tap of T2 and the bias voltage of the switch to the transistor. In this way, the gate voltage swing of the switching device is immune to the LO power. The mixer's stability can be improved by preventing changes in 1-dB compression power (P\u003csub\u003e1dB\u003c/sub\u003e) and CG by the power. The designed parameters of the proposed mixer are listed in Table\u0026nbsp;\u003cspan refid=\"Tab1\" class=\"InternalRef\"\u003e1\u003c/span\u003e.\u003c/p\u003e \u003cp\u003eThe mixer's conversion gain is a crucial parameter. The switching pair approximates an ideal switch when the LO signal is an ideal square wave. Ref. [\u003cspan citationid=\"CR9\" class=\"CitationRef\"\u003e9\u003c/span\u003e] provides the voltage conversion gain of a typical mixer when the LO signal is an ideal sine wave.\u003cdiv id=\"Equ1\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ1\" name=\"EquationSource\"\u003e\n$${A_{V,conv}} \\approx \\frac{2}{\\pi }{g_m}{R_L}\\left[ {1 - \\frac{{\\sqrt 2 {{\\left( {VGS - VTH} \\right)}_{sw}}}}{{\\pi {V_{LO}}}}} \\right]$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e1\u003c/div\u003e\u003c/div\u003e\u003c/p\u003e \u003cp\u003ewhere \u003cem\u003e(V\u003c/em\u003e\u003csub\u003e\u003cem\u003eGS\u003c/em\u003e\u003c/sub\u003e\u003cem\u003e-V\u003c/em\u003e\u003csub\u003e\u003cem\u003eTH\u003c/em\u003e\u003c/sub\u003e\u003cem\u003e)\u003c/em\u003e\u003csub\u003e\u003cem\u003esw\u003c/em\u003e\u003c/sub\u003e the switching tube's overdrive voltage, \u003cem\u003eR\u003c/em\u003e\u003csub\u003e\u003cem\u003eL\u003c/em\u003e\u003c/sub\u003e the mixer load impedance, \u003cem\u003eV\u003c/em\u003e\u003csub\u003e\u003cem\u003eLO\u003c/em\u003e\u003c/sub\u003e the magnitude of the local oscillation signal, and \u003cem\u003eg\u003c/em\u003e\u003csub\u003e\u003cem\u003em\u003c/em\u003e\u003c/sub\u003e the RF input stage's trans-conductance.\u003c/p\u003e \u003cp\u003eIn order to get the desired voltage gain of the mixer, increasing \u003cem\u003eg\u003c/em\u003e\u003csub\u003e\u003cem\u003em\u003c/em\u003e\u003c/sub\u003e would be an effective way. However, high \u003cem\u003eg\u003c/em\u003e\u003csub\u003e\u003cem\u003em\u003c/em\u003e\u003c/sub\u003e would increase the circuit's power consumption, and boost the load impedance \u003cem\u003eR\u003c/em\u003e\u003csub\u003e\u003cem\u003eL\u003c/em\u003e\u003c/sub\u003e, thus limiting the CG of the circuit. The \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP\u003c/sub\u003e charging and discharging currents of the common-source nodes P and Q appear as spike pulses when the LO signal amplitude is large. A strong LO signal causes the high leakage current. If the overdrive voltage of the switch device is reduced, the size of the switch device is increased to remain the same the current flowing through the device, resulting in a reduction in the mixer's performance. To keep the device be operated by LO signal, the transistor is biased near the edge of the NMOS triode area, i.e., \u003cem\u003eV\u003c/em\u003e\u003csub\u003e\u003cem\u003eGS\u003c/em\u003e\u003c/sub\u003e \u003cem\u003e= V\u003c/em\u003e\u003csub\u003e\u003cem\u003eTH\u003c/em\u003e\u003c/sub\u003e. The MOS device may reach a narrower gate width \u003cem\u003eW\u003c/em\u003e because of the gain-boosting and noise-canceling techniques utilized in the trans-conductance stage construction. This can decrease parasitic capacitance and lower the overdrive voltage, which results in a reduction in voltage gain.\u003c/p\u003e \u003cp\u003e \u003cdiv class=\"gridtable\"\u003e\u003ctable float=\"Yes\" id=\"Tab1\" border=\"1\"\u003e \u003ccaption language=\"En\"\u003e \u003cdiv class=\"CaptionNumber\"\u003eTable 1\u003c/div\u003e \u003cdiv class=\"CaptionContent\"\u003e \u003cp\u003eComponent parameters of the proposed mixer.\u003c/p\u003e \u003c/div\u003e \u003c/caption\u003e \u003ccolgroup cols=\"4\"\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c1\" colnum=\"1\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c2\" colnum=\"2\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c3\" colnum=\"3\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c4\" colnum=\"4\"\u003e\u003c/div\u003e \u003cthead\u003e \u003ctr\u003e \u003cth align=\"left\" colname=\"c1\"\u003e \u003cp\u003eDevice\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colname=\"c2\"\u003e \u003cp\u003eParameter\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colname=\"c3\"\u003e \u003cp\u003eDevice\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colname=\"c4\"\u003e \u003cp\u003eParameter\u003c/p\u003e \u003c/th\u003e \u003c/tr\u003e \u003c/thead\u003e \u003ctbody\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e1,4\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e20\u0026micro;m /0.1\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eR\u003c/em\u003e\u003csub\u003e\u003cem\u003e1\u003c/em\u003e\u003c/sub\u003e\u003cem\u003e/R\u003c/em\u003e\u003csub\u003e\u003cem\u003e3\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e2 kΩ\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e2,5\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e20\u0026micro;m /0.1\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eR\u003c/em\u003e\u003csub\u003e\u003cem\u003e2\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e100 Ω\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e3,6\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e100\u0026micro;m /0.1\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eR\u003c/em\u003e\u003csub\u003e\u003cem\u003e4\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e40 kΩ\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e7,10\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e46\u0026micro;m /0.1\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e100 pH\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e11,12\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e60\u0026micro;m /0.2\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e70 pH\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e13\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e120\u0026micro;m /0.2\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eR\u003c/em\u003e\u003csub\u003e\u003cem\u003eL\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e500 Ω\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e14,15\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e30\u0026micro;m /0.1\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eV\u003c/em\u003e\u003csub\u003e\u003cem\u003eb1\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e0.7 V\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e16,17\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e96\u0026micro;m /0.1\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eV\u003c/em\u003e\u003csub\u003e\u003cem\u003eb2\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e0.85 V\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e(\u003cem\u003eW/L\u003c/em\u003e)\u003csub\u003e18,19\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e160\u0026micro;m /0.1\u0026micro;m\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e\u003cem\u003eV\u003c/em\u003e\u003csub\u003e\u003cem\u003eb3\u003c/em\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e1 V\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003c/tbody\u003e \u003c/colgroup\u003e \u003c/table\u003e\u003c/div\u003e \u003c/p\u003e \u003cp\u003eThe suggested mixer, as shown in Fig.\u0026nbsp;3, employs an enhanced gain-boosting circuit based on the g\u003csub\u003em\u003c/sub\u003e-boosting approach and the noise-canceling technique utilized in the LNA in Ref. [\u003cspan citationid=\"CR10\" class=\"CitationRef\"\u003e10\u003c/span\u003e]. Transistors M\u003csub\u003e1\u003c/sub\u003e-M\u003csub\u003e3\u003c/sub\u003e are used to convert the RF input voltage into RF current. During the first stage, the common-gate transistor is an input-matched network with an impedance of 1/g\u003csub\u003em\u003c/sub\u003e. The input impedance of the trans-conductance stage is comparable to the input impedance of the common-gate amplifier when the noise caused by the common gate transistor is considered. A high input impedance and equivalent trans-conductance may be attained.\u003cdiv id=\"Equ2\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ2\" name=\"EquationSource\"\u003e\n$$Zin=\\frac{{rO1}}{{1+\\left( {gm1+gmb1} \\right)rO1}}$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e2\u003c/div\u003e\u003c/div\u003e\u003cdiv id=\"Equ3\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ3\" name=\"EquationSource\"\u003e\n$$Gm \\approx gm1gm2rO1+\\frac{{gm3}}{{1+gm3R2+{{R2} \\mathord{\\left/ {\\vphantom {{R2} {R3}}} \\right. \\kern-0pt} {R3}}}}$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e3\u003c/div\u003e\u003c/div\u003e\u003c/p\u003e \u003cp\u003eM\u003csub\u003e3\u003c/sub\u003e is a common-source transistor with negative feedback at the source stage, allowing the output device to become more linear in order to make up for the mixer design's IIP3 performance. The current from the LO stage is utilized by M\u003csub\u003e3\u003c/sub\u003e as the trans-conductance current when the transistor's gate is connected with the RF input. Its gain is increased due to the high output impedance of the common-gate stage matching network. In addition, in order to increase the equivalent trans-conductance of the circuit and suppress the noise, a g\u003csub\u003em\u003c/sub\u003e -boosting circuit is added after the matching circuit. The gm-boosting circuit consists of two amplifiers (M\u003csub\u003e2\u003c/sub\u003e, M\u003csub\u003e3\u003c/sub\u003e) in a common-source configuration. The topology has a higher gain compared to the common-source amplifier by adding the input signal to the gate of transistor M\u003csub\u003e3\u003c/sub\u003e.\u003cdiv id=\"Equ4\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ4\" name=\"EquationSource\"\u003e\n$$Zout=\\left[ {\\left( {1+gm3rO3} \\right)R2+rO3} \\right]\\parallel rO2$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e4\u003c/div\u003e\u003c/div\u003e\u003cdiv id=\"Equ5\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ5\" name=\"EquationSource\"\u003e\n$$AV,pro \\approx \\left[ {gm1gm2rO1+\\frac{{gm3}}{{1+gm3R2+{{R2} \\mathord{\\left/ {\\vphantom {{R2} {R3}}} \\right. \\kern-0pt} {R3}}}}} \\right]\\cdot \\left[ {\\left( {gm3rO3R2} \\right)\\parallel rO2} \\right]$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e5\u003c/div\u003e\u003c/div\u003e\u003c/p\u003e \u003cp\u003eTo evaluate the sizes of M\u003csub\u003e2\u003c/sub\u003e and M\u003csub\u003e3\u003c/sub\u003e, high g\u003csub\u003em\u003c/sub\u003e is necessary as it is proportional to the gain of the mixer, which can reduce the NF. So large M\u003csub\u003e3\u003c/sub\u003e is always required. However, big size could lead to large gate-source capacitance (\u003cem\u003eC\u003c/em\u003e\u003csub\u003eGS\u003c/sub\u003e) and gate-drain capacitance (\u003cem\u003eC\u003c/em\u003e\u003csub\u003eGD\u003c/sub\u003e), which could impact the circuit's input matching and increase the noise. Therefore, a PMOS dynamic current switching circuit with a tuned inductor is adopted in the design in the paper.\u003c/p\u003e \u003cp\u003eWhen the LO signal is positive, the switch turns on transistors M\u003csub\u003e7\u003c/sub\u003e and M\u003csub\u003e9\u003c/sub\u003e, while turns off transistors M\u003csub\u003e8\u003c/sub\u003e and M\u003csub\u003e10\u003c/sub\u003e. During the second half cycle, the switches are opposite. During the switching, the varying \u003cem\u003ev\u003c/em\u003e\u003csub\u003en\u003c/sub\u003e modulates the switching time, and the \u003cem\u003ev\u003c/em\u003e\u003csub\u003en\u003c/sub\u003e moves directly to the IF output to affect the NF of the circuit. As seen in Fig.\u0026nbsp;4, noise can be generated when both switching devices are turned on simultaneously, or at the zero-crossing point.\u003c/p\u003e \u003cp\u003eThe switching noise is the main part of the mixer's whole noise performance. The output resistance can be increased by decreasing the bias current through the switching devices. This can lower the switching device's overdrive voltage without affecting the bias current through the trans-conductance stage. Dynamic Current-Bleeding (DCB) [\u003cspan citationid=\"CR11\" class=\"CitationRef\"\u003e11\u003c/span\u003e] is a method that can greatly improve noise performance as well as mixer gain performance. In the DCB approach, certain current is required to be injected at the moment of switching pair switching to eliminate the flicker noise at the output.\u003c/p\u003e \u003cp\u003eDue to lower flicker noise angle of PMOS device, the PMOS based switching circuit can prevent the noise injection. It can sharpen the on/off of the LO switching stage by injecting current into the mixer core during the on/off switching. As shown in Fig.\u0026nbsp;5a, a similar fixed current injection is used in the traditional static current injection approach. It is ineffective for reducing the direct flicker noise since it creates more white noise and increases the effect of parasitic capacitance at the common-source node. If a dynamic current is injected at the common-source node, as shown in Fig.\u0026nbsp;5b, the problems encountered in injecting a fixed current can be avoided. However, there is still the influence of device capacitance on the noise of the circuit in Fig.\u0026nbsp;5b. To further enhance the NF of the PMOS switch current injection circuit, an efficient current injection technique is needed to lessen the injection of noise at all stages during the simultaneous conduction of the switching pair. In order to address these limitations and restrictions, a PMOS dynamic current injection approach with a tuned inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e is proposed, as shown in Fig.\u0026nbsp;5c.\u003c/p\u003e \u003cp\u003eIn Fig.\u0026nbsp;5c, a tuned inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e and a cross-coupled transistor pair (M\u003csub\u003e13\u003c/sub\u003e-M\u003csub\u003e15\u003c/sub\u003e) made of PMOS transistors are constructed. The voltages at the P and Q nodes are varied with the input LO signal, which are controlled by M\u003csub\u003e14\u003c/sub\u003e and M\u003csub\u003e15\u003c/sub\u003e when the current \u003cem\u003eI\u003c/em\u003e is injected into the common-source node. The branch current of the mixer switching pair is decreased by injecting current without altering the branch current of the trans-conductance stage during LO signal oscillation transition, so minimizing the noise pulses at the mixer's output IF port. To make sure that only the dynamic current can be injected with the LO signal at the zero-crossing point, a bias voltage design is needed. The bias voltage is small at the zero-crossing point. When the cross-coupled PMOS pair is injected, the top current source is turned on with current \u003cem\u003eI\u003c/em\u003e flowing into the common-source node. Away from the zero-crossing point, due to the higher bias voltage, the PMOS device is disconnected. So there isn\u0026rsquo;t any injected current and the mixer works normally. With this method, the mixer's flicker noise is significantly reduced without adding more white noise. The mixer's linearity and bandwidth performance are not deteriorated at all. M\u003csub\u003e14\u003c/sub\u003e and M\u003csub\u003e15\u003c/sub\u003e need to be sized reasonably to inject enough current while avoiding extra parasitic capacitance to the switching pair's common-source end. In particular, the parasitic effect might produce a different flicker noise. Therefore, the tuned inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e is added in the PMOS cross-coupling pair to balance the PMOS device's device capacitance and enhance the mixer circuit's NF. According to the final testing findings, the circuit with the inductor shows 3 dB better NF than the other ones.\u003c/p\u003e \u003cp\u003eFigure 6 shows the effects of the parasitic capacitors on the device. \u003cem\u003eC\u003c/em\u003e\u003csub\u003eGS\u003c/sub\u003e in the saturation zone can influence the frequency response. The inter-stage parasitic capacitance \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP\u003c/sub\u003e can bypass some signal current to ground, which is another factor contributing to the considerable deterioration of the common-source common-gate amplifier in the millimeter-wave region. The high-frequency operating characteristics of the mixer are substantially controlled by the parasitic capacitance at the tail nodes P and Q of the switch pair, as illustrated in Fig.\u0026nbsp;2, which also has a considerable influence on the mixer's NF and gain.\u003c/p\u003e \u003cp\u003eFigure 7a shows typical method for removing parasitic capacitance by adding an inter-stage series inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eS\u003c/em\u003e\u003c/sub\u003e. A π-shaped matching network is created by \u003cem\u003eL\u003c/em\u003e\u003csub\u003eS\u003c/sub\u003e with the parasitic capacitors \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP1\u003c/sub\u003e and \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP2\u003c/sub\u003e. In fact, since the difference between the common-source output resistance \u003cem\u003er\u003c/em\u003e\u003csub\u003e\u003cem\u003eO3\u003c/em\u003e\u003c/sub\u003e and the common-gate input resistance 1/g\u003csub\u003em\u003c/sub\u003e is so considerable, it is difficult to match with only one inductor and two parasitic capacitors. Employing two series inductors for differential circuit design would need extra chip area.\u003c/p\u003e \u003cp\u003eA series-parallel inter-stage hybrid LC network is shown in Fig.\u0026nbsp;7b. The signal current shunted by \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP2\u003c/sub\u003e is reduced by resonating \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e with \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP2\u003c/sub\u003e. The source of M\u003csub\u003e7\u003c/sub\u003e has a tail node impedance \u003cem\u003eZ\u003c/em\u003e\u003csub\u003eP\u003c/sub\u003e of:\u003cdiv id=\"Equ6\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ6\" name=\"EquationSource\"\u003e\n$$Zp=rO3\\parallel sLP\\parallel \\frac{1}{{j\\omega LOCP}}$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e6\u003c/div\u003e\u003c/div\u003e\u003c/p\u003e \u003cp\u003ewhere \u003cem\u003er\u003c/em\u003e\u003csub\u003e\u003cem\u003eO3\u003c/em\u003e\u003c/sub\u003e represents the small signal output resistance of transistor M\u003csub\u003e3\u003c/sub\u003e.\u003c/p\u003e \u003cp\u003eSince the LO signal drives the switching device set in the saturation operation region, its \u003cem\u003eZ\u003c/em\u003e\u003csub\u003eP\u003c/sub\u003e is quite small. Practically, resonance with \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP2\u003c/sub\u003e requires a quite big inductor with large area. In addition, the PMOS\u0026rsquo; nonlinear capacitance tends to produce harmonics and create leakage for the insertion current with the increment of the PMOS size.\u003c/p\u003e \u003cp\u003eAn improved LC resonant network with the resonant point at the LO frequency of 77.2 GHz is designed, as shown in Fig.\u0026nbsp;7c. A peak inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e at the common-source end of the LO switching devices is added to improve the noise performance. It can reduce the flicker noise by canceling the influence of \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP2\u003c/sub\u003e between the switching and trans-conductance stages, which also increases the CG due to the lack of signal leakage through the parasitic capacitance. The parallel inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e is the tuned inductor in the dynamic current injection module. The LC parallel resonant network with the inter-stage parasitic capacitor \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP1\u003c/sub\u003e can neutralize the parasitic capacitance of the PMOS cross-coupling to the circuit. The tuned circuit can suppress the harmonics and the leakage current caused by the inductive filling effect in the nonlinear capacitance. The RF high-impedance circuit by the inductor and \u003cem\u003eC\u003c/em\u003e\u003csub\u003eP\u003c/sub\u003e at circuit nodes P and Q can decrease the loss of the switching circuit near the RF frequency. The indirect formation of flicker noise by the LO big signal charging and discharging low resistance channel can be reduced due to the closer frequency of the RF and LO signals.\u003c/p\u003e \u003cp\u003eThe proposed mixer circuit in the paper is assisted by the \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e and \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e resonance effects, with \u003cem\u003eZ\u003c/em\u003e\u003csub\u003eP\u003c/sub\u003e being approximated by \u003cem\u003er\u003c/em\u003e\u003csub\u003e\u003cem\u003eO3\u003c/em\u003e\u003c/sub\u003e. The higher impedance of the switching stage corresponds to better common mode interference rejection at the RF input port. The inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e needs to be optimized for the design, and its value is chosen by maximum linearity and minimum NF. The peak gain of the mixer increases by roughly 12 dB and the noise decreases by around 9 dB when the inter-stage parallel inductors \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e and \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e are added.\u003c/p\u003e \u003cp\u003ePMOS cross pairs are utilized as active loads to reduce the load-level noise. The system noise is increased by the current injection module of the circuit. The cross-coupling approach is used to mitigate the noise deterioration. The gate length \u003cem\u003eL\u003c/em\u003e of the MOS device can be chosen as large as necessary to reduce the flicker noise of the transistor and to lessen its channel modulation impact, in favor of increasing the output impedance and the gain of the mixer.\u003c/p\u003e \u003cp\u003eAs long as the buffer circuit itself does not introduce excessive signal attenuation, no additional noise can be generated due to the high gain of the mixer before the IF buffer circuit. A self-biasing inverting amplifier is used as the IF output buffer circuit, including M\u003csub\u003e10\u003c/sub\u003e-M\u003csub\u003e13\u003c/sub\u003e and \u003cem\u003eR\u003c/em\u003e\u003csub\u003e4\u003c/sub\u003e. The load circuit\u0026rsquo;s output signal swing is considerable, and the buffer needs a sizable enough linear range to guard the mixer\u0026rsquo;s linearity.\u003c/p\u003e \u003cp\u003eThe input and output DC levels can be expressed as:\u003cdiv id=\"Equ7\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ7\" name=\"EquationSource\"\u003e\n$$Vin=Vout=\\frac{{(VDD+VTHP+VTHN)}}{2}$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e7\u003c/div\u003e\u003c/div\u003e\u003c/p\u003e \u003cp\u003eThe transistor's stability factor is defined to be K\u0026thinsp;\u0026gt;\u0026thinsp;1 over the whole frequency range. It is feasible to improve the output gain by adjusting the value of \u003cem\u003eR\u003c/em\u003e\u003csub\u003e4\u003c/sub\u003e. To keep the amplifier's front stage from being impacted by the input signal's DC level, the off-chip blocking capacitors \u003cem\u003eC\u003c/em\u003e\u003csub\u003e4\u003c/sub\u003e is used. The buffer circuit can reduce the complexity of the circuit since no additional bias circuit is needed. The flicker noise and the channel modulation effect can be minimized by increasing the gate length \u003cem\u003eL\u003c/em\u003e and gate width \u003cem\u003eW\u003c/em\u003e of MOS devices. Filtering is necessary to obtain the IF signal because fundamental signals and harmonic signals can leak from the RF and LO ports to the IF side. Capacitor filtering is employed since each harmonic signal's frequency is far from that of the zero IF signal.\u003c/p\u003e"},{"header":"3 Simulation Results","content":"\u003cp\u003eThe mixer proposed in this research is implemented in TSMC 90-nm RF CMOS process and simulated with ADS (Advanced Design System). Figure\u0026nbsp;8a shows the way the overall circuit of the mixer is implemented. The RF and LO ports convert single-ended to differential signals using an external ideal transformer during simulation. Both the trans-conductance stage of the mixer and the LO switching stage employ the same load resistance and same bias for a fair comparison. Figure\u0026nbsp;8b shows a micrograph of the circuit with a chip area of 0.5 \u0026times; 0.5mm\u003csup\u003e2\u003c/sup\u003e for the core circuit of the mixer.\u003c/p\u003e \u003cp\u003eThe simulation results for the mixer with various inter-stage network architectures are compiled in Fig.\u0026nbsp;9a and Fig.\u0026nbsp;9b, respectively. Figure\u0026nbsp;9a shows the simulation results of the LO input power versus the conversion gain. With RF frequencies 71\u0026ndash;86 GHz and IF frequencies at 200 MHz, the mixer's CG varies from 12.6 dB to 18.4 dB. The maximum CG is achieved when LO power is 2 dBm. The PMOS dynamic current injection can increase the mixer's voltage gain. The mixer shows CG of 18.4 dB due to the addition of the inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e and the improved gain-boosting circuit in the trans-conductance stage.\u003c/p\u003e \u003cp\u003eAs shown in Fig.\u0026nbsp;9b, the addition of the inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eP\u003c/em\u003e\u003c/sub\u003e lowers the mixer's SSB NF by 2.6 dB. The SSB NF is decreased by around 3.3 dB by adopting a dynamic current injection circuit with tuned inductor \u003cem\u003eL\u003c/em\u003e\u003csub\u003e\u003cem\u003eT\u003c/em\u003e\u003c/sub\u003e, with a final SSB NF of 12.2 dB.\u003c/p\u003e \u003cp\u003e \u003c/p\u003e\u003cp\u003eAs shown in Fig.\u0026nbsp;10a, with the frequency from 71 GHz to 86 GHz, the reflection coefficient \u003cem\u003eS\u003c/em\u003e\u003csub\u003e11\u003c/sub\u003e is less than \u0026minus;\u0026thinsp;40 dB at the RF port, and the transmission coefficient \u003cem\u003eS\u003c/em\u003e\u003csub\u003e21\u003c/sub\u003e is less than \u0026minus;\u0026thinsp;30 dB at the IF port. Figure\u0026nbsp;10b displays the test results for the LO-to-IF port, LO-to-RF port, and RF-to-IF port isolation, with isolation values larger than 50 dB, 60 dB, and 45 dB, respectively. The mixer also provides enough linearity to maintain high-gain and low-noise. The IP\u003csub\u003e1dB\u003c/sub\u003e and the IIP3 at RF frequency of 77 GHz are \u0026minus;\u0026thinsp;17.2 and \u0026minus;\u0026thinsp;6.14 dBm, respectively. Figure\u0026nbsp;11a shows simulated and measured conversion gain and NF versus LO power, with fixed LO of 77.2 GHz and RF of 77 GHz. Figure\u0026nbsp;11b shows the relationship between IF output power and RF input power. Figure\u0026nbsp;12a shows the relationship between RF input power and the conversion gain. Figure\u0026nbsp;12b shows the impact of various IF frequencies on the gain performance with 40 dBm RF power. The mixer\u0026rsquo;s core circuit consumes around 6 mA of current at a supply voltage of 1.2 V.\u003c/p\u003e \u003cp\u003e \u003c/p\u003e \u003cp\u003eFig. 12 (a) Simulation results of the RF input power versus the conversion gain. (b) Simulation results of conversion gain at different IF frequencies.\u003c/p\u003e \u003cp\u003eA figure-of-merit (FOM) suitable for evaluating the performance of a down-conversion mixer, a low-noise amplifier, or a receiver front-end can be defined as below [\u003cspan citationid=\"CR12\" class=\"CitationRef\"\u003e12\u003c/span\u003e]:\u003cdiv id=\"Equ8\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ8\" name=\"EquationSource\"\u003e\n$$FOM1=10\\log \\left( {\\frac{{{{10}^{CG[dB]/20}} \\cdot {{10}^{\\left( {IIP3[dBm] - 10} \\right)/20}}}}{{{{10}^{NF[dB]/10}} \\cdot PDC[mW]}} \\cdot \\frac{{fRF}}{{1GHz}}} \\right)$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e8\u003c/div\u003e\u003c/div\u003e\u003c/p\u003e \u003cp\u003ewhere \u003cem\u003eCG\u003c/em\u003e[dB] the average CG in magnitude; \u003cem\u003eIIP3\u003c/em\u003e[dBm] represents the input third-order intercept point, \u003cem\u003eNF\u003c/em\u003e[dB] the DSB NF in magnitude, \u003cem\u003eP\u003c/em\u003e\u003csub\u003eDC\u003c/sub\u003e[mW] power dissipation in milliwatts, \u003cem\u003ef\u003c/em\u003e\u003csub\u003e\u003cem\u003eRF\u003c/em\u003e\u003c/sub\u003e the value of the RF input frequency.\u003c/p\u003e \u003cp\u003eThe FOM shown in Eq.\u0026nbsp;\u003cspan refid=\"Equ8\" class=\"InternalRef\"\u003e8\u003c/span\u003e includes the most relevant parameters for evaluating down-conversion mixer for low-power, high-gain, low-noise, and high-linearity applications. Moreover, in numerous pieces of previous work, the information of NF and linearity are not available. To compare with them, the following simplified FOM can be used.\u003cdiv id=\"Equ9\" class=\"Equation\"\u003e\u003cdiv format=\"TEX\" class=\"mathdisplay\" id=\"FileID_Equ9\" name=\"EquationSource\"\u003e\n$$FOM2=10\\log \\left( {\\frac{{{{10}^{CG[dB]/20}}}}{{PDC[mW]}} \\cdot \\frac{{fRF}}{{1GHz}}} \\right)$$\u003c/div\u003e\u003cdiv class=\"EquationNumber\"\u003e9\u003c/div\u003e\u003c/div\u003e\u003c/p\u003e \u003cp\u003eTable\u0026nbsp;\u003cspan refid=\"Tab2\" class=\"InternalRef\"\u003e2\u003c/span\u003e summarizes the performance of the proposed down-conversion mixer and compares the parameters with the recently reported mixers. Due to the application of an improved gain-boosting approach, the proposed mixer exhibits a significant enhancement in CG and the noise compared with the mixers published in [\u003cspan citationid=\"CR13\" class=\"CitationRef\"\u003e13\u003c/span\u003e], [\u003cspan citationid=\"CR14\" class=\"CitationRef\"\u003e14\u003c/span\u003e]. Comparing with the 76\u0026ndash;77 GHz CMOS mixer in [\u003cspan citationid=\"CR9\" class=\"CitationRef\"\u003e9\u003c/span\u003e], the proposed mixer shows better CG, larger port-to-port isolation, and larger FOM value. The proposed mixer consumes less power than the mixers described in [\u003cspan citationid=\"CR9\" class=\"CitationRef\"\u003e9\u003c/span\u003e], [\u003cspan citationid=\"CR14\" class=\"CitationRef\"\u003e14\u003c/span\u003e], and [\u003cspan citationid=\"CR15\" class=\"CitationRef\"\u003e15\u003c/span\u003e]. In comparison to the mixer described in [\u003cspan citationid=\"CR16\" class=\"CitationRef\"\u003e16\u003c/span\u003e], the proposed mixer offers better noise performance. It can be concluded that the proposed mixer exhibits a competitive performance with the low-power, the high-gain and the low-noise performance.\u003c/p\u003e"},{"header":"4 Conclusions","content":"\u003cp\u003eIn the paper, a novel mixer is designed, with the improved gain and better noise performance. In the design, an improved trans-conductance circuit and a PMOS dynamic current switching technique with inductance are adopted. At the input transistor's trans-conductance stage, the gain-boosting approach is used to increase the trans-conductance and the output resistance. At the same time, the circuit noise can be suppressed effectively. The second-order derivative trans-conductance is kept zero in the input transistor by using the optimal biasing technique, which enhances the linearity without decreasing the gain. To further increase the mixer\u0026rsquo;s gain and better NF, a dynamic current injection circuit with a tuned inductor is added at the LO input stage. Simulation results based on TSMC 90-nm CMOS process show that the proposed mixer consumes only 6.96 mW with 1.2 V supply, and DSB NF of 9.2 dB, CG of +\u0026thinsp;18.4 dB, and OIP3 of 12.2 dBm. The proposed mixer shows potential application in the field of vehicle radar system, providing the benefits of high-gain, low-noise, and low power consumption, \u003cem\u003eetc.\u003c/em\u003e.\u003c/p\u003e \u003cp\u003e \u003cdiv class=\"gridtable\"\u003e\u003ctable float=\"Yes\" id=\"Tab2\" border=\"1\"\u003e \u003ccaption language=\"En\"\u003e \u003cdiv class=\"CaptionNumber\"\u003eTable 2\u003c/div\u003e \u003cdiv class=\"CaptionContent\"\u003e \u003cp\u003eSummary results and performance comparison of other mixers.\u003c/p\u003e \u003c/div\u003e \u003c/caption\u003e \u003ccolgroup cols=\"7\"\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c1\" colnum=\"1\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c2\" colnum=\"2\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c3\" colnum=\"3\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c4\" colnum=\"4\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c5\" colnum=\"5\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c6\" colnum=\"6\"\u003e\u003c/div\u003e \u003cdiv align=\"left\" class=\"colspec\" colname=\"c7\" colnum=\"7\"\u003e\u003c/div\u003e \u003cthead\u003e \u003ctr\u003e \u003cth align=\"left\" colname=\"c1\" morerows=\"1\" rowspan=\"2\"\u003e \u003cp\u003ePerformance parameter\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colspan=\"6\" nameend=\"c7\" namest=\"c2\"\u003e \u003cp\u003eReferences\u003c/p\u003e \u003c/th\u003e \u003c/tr\u003e \u003ctr\u003e \u003cth align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003eThis work\u003c/b\u003e\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colname=\"c3\"\u003e \u003cp\u003e[\u003cspan citationid=\"CR9\" class=\"CitationRef\"\u003e9\u003c/span\u003e]\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colname=\"c4\"\u003e \u003cp\u003e[\u003cspan citationid=\"CR13\" class=\"CitationRef\"\u003e13\u003c/span\u003e]\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colname=\"c5\"\u003e \u003cp\u003e[\u003cspan citationid=\"CR14\" class=\"CitationRef\"\u003e14\u003c/span\u003e]\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colname=\"c6\"\u003e \u003cp\u003e[\u003cspan citationid=\"CR15\" class=\"CitationRef\"\u003e15\u003c/span\u003e]\u003c/p\u003e \u003c/th\u003e \u003cth align=\"left\" colname=\"c7\"\u003e \u003cp\u003e[\u003cspan citationid=\"CR16\" class=\"CitationRef\"\u003e16\u003c/span\u003e]\u003c/p\u003e \u003c/th\u003e \u003c/tr\u003e \u003c/thead\u003e \u003ctbody\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cb\u003eTechnology\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e90-nm\u003c/b\u003e\u003c/p\u003e \u003cp\u003e\u003cb\u003eCMOS\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e65-nm\u003c/p\u003e \u003cp\u003eCMOS\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e90-nm\u003c/p\u003e \u003cp\u003eCMOS\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e90-nm\u003c/p\u003e \u003cp\u003eCMOS\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e0.18-\u0026micro;m\u003c/p\u003e \u003cp\u003eCMOS\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e90-nm\u003c/p\u003e \u003cp\u003eCMOS\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cb\u003eFreq.\u0026nbsp;(GHz)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e71\u0026ndash;86\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e75\u0026ndash;90\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e70\u0026ndash;88\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e75\u0026ndash;85\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e62\u0026ndash;85\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e72\u0026ndash;100\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eCG\u003c/span\u003e \u003cb\u003e(dB)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e18.4\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e5\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e5.3-9\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e1.5\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e-4.3\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e17.1\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eNF\u003c/span\u003e \u003cb\u003e(dB)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e9.2\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003eN/A\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e22\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e23.3\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003eN/A\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e16.4\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eIP\u003c/span\u003e\u003csub\u003e\u003cb\u003e1dB\u003c/b\u003e\u003c/sub\u003e \u003cb\u003e(dBm)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e-17.2\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e2.5\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e-13\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e-9\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e2.14\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e-13\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eOIP3\u003c/span\u003e \u003cb\u003e(dBm)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e12.23\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003eN/A\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003eN/A\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003eN/A\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003eN/A\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e14.1\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eP\u003c/span\u003e\u003csub\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eLO\u003c/span\u003e\u003c/sub\u003e \u003cb\u003e(dBm)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e2\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e0\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e-4\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e5\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e5\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e0\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cb\u003eISO. (dB)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e\u0026gt;\u0026thinsp;45\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e37\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e\u0026gt;\u0026thinsp;40 \u003csup\u003ea\u003c/sup\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e49.2\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e30\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e43.3\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cb\u003eSize (mm\u003c/b\u003e\u003csup\u003e\u003cb\u003e2\u003c/b\u003e\u003c/sup\u003e\u003cb\u003e)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e0.25\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e0.14\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e0.32\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e0.62\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e0.38\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e0.61\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eV\u003c/span\u003e\u003csub\u003e\u003cb\u003eDD\u003c/b\u003e\u003c/sub\u003e \u003cb\u003e(V)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e1.2\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e1.5\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e1.2\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e1.2\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e1.2\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e1.2\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eP\u003c/span\u003e\u003csub\u003e\u003cb\u003eDC\u003c/b\u003e\u003c/sub\u003e \u003cb\u003e(mW)\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e6.96\u003c/b\u003e \u003csup\u003e\u003cb\u003eb\u003c/b\u003e\u003c/sup\u003e \u003cb\u003e/10.56\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e8 \u003csup\u003eb\u003c/sup\u003e/12\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e5\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e13\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e10.8\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e8.6\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eFOM\u003c/span\u003e\u003csub\u003e\u003cb\u003e1\u003c/b\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e2.36\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003eN/A\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e-11.94\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e-18.3\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003eN/A\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e-4.98\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003ctr\u003e \u003ctd align=\"left\" colname=\"c1\"\u003e \u003cp\u003e\u003cspan type=\"BoldItalic\" class=\"BoldItalic\" name=\"Emphasis\"\u003eFOM\u003c/span\u003e\u003csub\u003e\u003cb\u003e2\u003c/b\u003e\u003c/sub\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c2\"\u003e \u003cp\u003e\u003cb\u003e19.64\u003c/b\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c3\"\u003e \u003cp\u003e12.28\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c4\"\u003e \u003cp\u003e15.5\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c5\"\u003e \u003cp\u003e9.4 \u003csup\u003ec\u003c/sup\u003e\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c6\"\u003e \u003cp\u003e8.6\u003c/p\u003e \u003c/td\u003e \u003ctd align=\"left\" colname=\"c7\"\u003e \u003cp\u003e18.1\u003c/p\u003e \u003c/td\u003e \u003c/tr\u003e \u003c/tbody\u003e \u003c/colgroup\u003e \u003ctfoot\u003e \u003ctr\u003e\u003ctd colspan=\"7\"\u003e\u003csup\u003ea\u003c/sup\u003e 2LO- to-RF Isolation \u003csup\u003eb\u003c/sup\u003e Without buffer \u003csup\u003ec\u003c/sup\u003e At 80 GHz\u003c/td\u003e\u003c/tr\u003e \u003c/tfoot\u003e \u003c/table\u003e\u003c/div\u003e \u003c/p\u003e"},{"header":"Declarations","content":"\u003cp\u003e\u003cstrong\u003eEthics Approval and Consent to Participate\u003c/strong\u003e Not applicable.\u003c/p\u003e\n\u003cp\u003e\u003cstrong\u003eCompeting interests\u003c/strong\u003e The authors declare no competing interests.\u003c/p\u003e\n\u003cp\u003e\u003cstrong\u003eAuthors\u0026apos; contributions\u003c/strong\u003e Huan Zhang and Yanfeng Jiang wrote the main manuscript text; Sida Tang and Mengye Cai assisted with measurements. All authors reviewed the manuscript and contributed with comments.\u003c/p\u003e\n\u003cp\u003e\u003cstrong\u003eFunding\u0026nbsp;\u003c/strong\u003eOpen access funding provided by The Natural Science Foundation of China with No.61774078.\u0026nbsp;\u003c/p\u003e\n\u003cp\u003e\u003cstrong\u003eAvailability of data and materials\u003c/strong\u003e The data that support the findings of this study are available from the corresponding author upon reasonable request.\u003c/p\u003e"},{"header":"References","content":"\u003col\u003e\n\u003cli\u003eYu J, Liao CH, Hsieh CH, Hu R, Niu DC: 77-110GHz 90nm-CMOS Receiver Design. \u003cem\u003e2013 Asia-Pacific Microwave Conference Proceedings (APMC) IEEE \u003c/em\u003e2013\u003cstrong\u003e:\u003c/strong\u003e233-235, https:// doi. org/ 10.1109/ apmc.2013.6695104.\u003c/li\u003e\n\u003cli\u003eLiao Y, Mei N, Zhang Z: Feedback followed bias generation scheme for complementary passive mixer. \u003cem\u003eElectronics Letters \u003c/em\u003e2018, 54\u003cstrong\u003e:\u003c/strong\u003e531-533, https:// doi. org/ 10.1049/ el.2017.4803.\u003c/li\u003e\n\u003cli\u003eTan GH, Ramiah H, Mak P-I, Martins RP: A 0.35V 520- uW 2.4-GHz Current-Bleeding Mixer With Inductive-Gate and Forward-Body Bias, Achieving \u0026gt;13-dB Conversion Gain and \u0026gt;55-dB Port-to-Port Isolation. \u003cem\u003eIEEE Transactions on Microwave Theory and Techniques \u003c/em\u003e2017, 65\u003cstrong\u003e:\u003c/strong\u003e1284-1293, https:// doi. org/ 10.1109/ TMTT.2016.2636143.\u003c/li\u003e\n\u003cli\u003ePark J, Lee C-H, Kim B-S, Laskar J: Design and Analysis of Low Flicker-Noise CMOS Mixers for Direct-Conversion Receivers. \u003cem\u003eIEEE Transactions on Microwave Theory and Techniques \u003c/em\u003e2006, 54\u003cstrong\u003e:\u003c/strong\u003e4372-4380, https:// doi. org/ 10.1109/ TMTT.2006.885582.\u003c/li\u003e\n\u003cli\u003eGuo B, Wang X, Chen H, Chen J: A 0.5\u0026ndash;6.5-GHz 3.9-dB NF 7.2-mW active down-conversion mixer in 65-nm CMOS. \u003cem\u003eModern Physics Letters B \u003c/em\u003e2018, 32, https:// doi. org/ 10.1142/ S0217984918502780.\u003c/li\u003e\n\u003cli\u003eRaheem MA, Hussain MMS: A Low Voltage NMOS current bleeding down conversion Mixer with source degeneration in 0.18\u0026micro;m CMOS Technology. In \u003cem\u003e2021 Devices for Integrated Circuit (DevIC)\u003c/em\u003e. pp. 637-6422021:637-642, https:// doi. org/ 10.1109/ DevIC50843.2021.9455921.\u003c/li\u003e\n\u003cli\u003eDarabi H, Chiu J: A noise cancellation technique in active RF-CMOS mixers. \u003cem\u003eIEEE Journal of Solid-State Circuits \u003c/em\u003e2005, 40\u003cstrong\u003e:\u003c/strong\u003e2628-2632, https:// doi. org/ 10.1109/ JSSC.2005.857428.\u003c/li\u003e\n\u003cli\u003eGuo B, Wang H, Chen J, Deilamsalehi MM: A CMOS low-noise active mixer with enhanced linearity and isolation by exploiting capacitive neutralization technique. \u003cem\u003eModern Physics Letters B \u003c/em\u003e2019, 33, https:// doi. org/ 10.1142/ S021798491950204X.\u003c/li\u003e\n\u003cli\u003ePan D, Duan Z, Huang L, Wang Y, Zhou Y, Wu B, Sun L: Design of high-linearity 75-90 GHz CMOS down-conversion mixer for automotive radar. \u003cem\u003eAnalog Integrated Circuits and Signal Processing \u003c/em\u003e2018, 97\u003cstrong\u003e:\u003c/strong\u003e313-322, https:// doi. org/ 10.1007/ s10470-018-1247-9.\u003c/li\u003e\n\u003cli\u003eChen WH, Gang L, Zdravko B, Niknejad AM: A Highly Linear Broadband CMOS LNA Employing Noise and Distortion Cancellation. \u003cem\u003eIEEE Journal of Solid-State Circuits \u003c/em\u003e2008, 43\u003cstrong\u003e:\u003c/strong\u003e1164-1176, https:// doi. org/ 10.1109/ JSSC.2008.920335.\u003c/li\u003e\n\u003cli\u003eLee J-Y, Yun T-Y: Low-Flicker-Noise and High-Gain Mixer Using a Dynamic Current-Bleeding Technique. \u003cem\u003eIEEE Microwave and Wireless Components Letters \u003c/em\u003e2017, 27\u003cstrong\u003e:\u003c/strong\u003e733-735, https:// doi. org/ 10.1109/ LMWC.2017.2723979.\u003c/li\u003e\n\u003cli\u003eSeyedi H, Dehdasht-Heydari R, Roshani S: UWB down-conversion mixer using an IM3 cancellation modified technique for zero and low IF applications. \u003cem\u003eMicroelectronics Journal \u003c/em\u003e2021, 109, https:// doi. org/ 10.1016/ j.mejo.2020.104983.\u003c/li\u003e\n\u003cli\u003eWu Y-C, Wang H: An E-band Double-Balanced Subharmonic Mixer With High Conversion Gain and Low Power in 90-nm CMOS Process. \u003cem\u003eIEEE Microwave and Wireless Components Letters \u003c/em\u003e2018, 28\u003cstrong\u003e:\u003c/strong\u003e70-72, https:// doi. org/ 10.1109/ LMWC.2017.2779847.\u003c/li\u003e\n\u003cli\u003eLin YS, Li GH: 13 mW 80 GHz down‐conversion mixer with 1.5 dB gain and 49.2 dB LO\u0026ndash;RF isolation. \u003cem\u003eElectronics Letters \u003c/em\u003e2014, 50\u003cstrong\u003e:\u003c/strong\u003e1449-1451, https:// doi. org/ 10.1049/ el.2014.1263.\u003c/li\u003e\n\u003cli\u003eChen Z, Liu H, Liu Z, Jiang Z, Yu Y, Wu Y, Zhao C, Kang K: A 62\u0026ndash;85-GHz High Linearity Upconversion Mixer With 18-GHz IF Bandwidth. \u003cem\u003eIEEE Microwave and Wireless Components Letters \u003c/em\u003e2019, 29\u003cstrong\u003e:\u003c/strong\u003e219-221, https:// doi. org/ 10.1109/ LMWC.2019.2894979.\u003c/li\u003e\n\u003cli\u003eLin Y-S, Lan K-S: A W-band CMOS down-conversion mixer using CMOS-inverter-based RF GM stage for conversion gain and linearity enhancement. \u003cem\u003eAnalog Integrated Circuits and Signal Processing \u003c/em\u003e2019, 99\u003cstrong\u003e:\u003c/strong\u003e133-146, https:// doi. org/ 10.1109/ RWS.2019.8714325.\u003c/li\u003e\n\u003c/ol\u003e"}],"fulltextSource":"","fullText":"","funders":[],"hasAdminPriorityOnWorkflow":false,"hasManuscriptDocX":true,"hasOptedInToPreprint":true,"hasPassedJournalQc":"","hasAnyPriority":false,"hideJournal":false,"highlight":"","institution":"","isAcceptedByJournal":true,"isAuthorSuppliedPdf":false,"isDeskRejected":"","isHiddenFromSearch":false,"isInQc":false,"isInWorkflow":false,"isPdf":false,"isPdfUpToDate":true,"isWithdrawnOrRetracted":false,"journal":{"display":true,"email":"[email protected]","identity":"journal-of-infrared-millimeter-and-terahertz-waves","isNatureJournal":false,"hasQc":true,"allowDirectSubmit":false,"externalIdentity":"ijim","sideBox":"Learn more about [Journal of Infrared, Millimeter, and Terahertz Waves](http://link.springer.com/journal/10762)","snPcode":"10762","submissionUrl":"https://submission.nature.com/new-submission/10762/3","title":"Journal of Infrared, Millimeter, and Terahertz Waves","twitterHandle":"","acdcEnabled":true,"dfaEnabled":true,"editorialSystem":"em","reportingPortfolio":"Springer Hybrid","inReviewEnabled":true,"inReviewRevisionsEnabled":false},"keywords":"77 GHz, down-conversion mixer, gain-boosting, LC resonant network","lastPublishedDoi":"10.21203/rs.3.rs-2373644/v1","lastPublishedDoiUrl":"https://doi.org/10.21203/rs.3.rs-2373644/v1","license":{"name":"CC BY 4.0","url":"https://creativecommons.org/licenses/by/4.0/"},"manuscriptAbstract":"\u003cp\u003eIn radio-frequency (RF) transceiver system, the receiver is used to convert RF signal to low/medium frequency for signal processing and information gathering. In the paper, a novel high-gain, low-noise CMOS mixer is designed and analyzed. In the design, gain-boosting and PMOS dynamic switching current are employed to achieve better effects of noise cancellation and trans-conductance enhancement. In order to improve the mixer's performance, the designed structure is mainly based on the double-balanced Gilbert lattice mixer combined with the parallel LC resonant and optimum biasing networks. The mixer shows high conversion gain (CG), low noise figure (NF), and low power consumption performances based on made possible by 90-nm CMOS technology. Operated at 77 GHz, the input third-order intercept point (IIP3) is -6.14 dBm, and the maximum conversion gain is 18.4 dB. At an IF frequency of 200 MHz, a bilateral band noise figure of 9.2 dB is recorded, while 6.96 mW is consumed with 1.2 V and 2 dBm LO power.\u003c/p\u003e \u003cp\u003e \u003c/p\u003e","manuscriptTitle":"77GHz Down-Conversion Mixer with +18.4 dB High-Gain, +12.2 dBm OIP3, and Low-Noise in 90-nm CMOS Technology","msid":"","msnumber":"","nonDraftVersions":[{"code":1,"date":"2022-12-19 15:34:14","doi":"10.21203/rs.3.rs-2373644/v1","editorialEvents":[{"type":"communityComments","content":0},{"type":"decision","content":"Major revision","date":"2023-03-04T09:17:43+00:00","index":"","fulltext":""},{"type":"editorInvitedReview","content":"","date":"2023-01-15T11:53:39+00:00","index":"hide","fulltext":""},{"type":"reviewerAgreed","content":"42be9ac5-b960-4de9-a415-6379b29bbb16","date":"2022-12-27T14:36:01+00:00","index":"hide","fulltext":""},{"type":"reviewersInvited","content":"","date":"2022-12-27T11:16:47+00:00","index":"","fulltext":""},{"type":"editorAssigned","content":"","date":"2022-12-26T22:27:03+00:00","index":"","fulltext":""},{"type":"checksComplete","content":"","date":"2022-12-15T09:05:40+00:00","index":"","fulltext":""},{"type":"submitted","content":"Journal of Infrared, Millimeter, and Terahertz Waves","date":"2022-12-13T11:05:47+00:00","index":"","fulltext":""}],"status":"published","journal":{"display":true,"email":"[email protected]","identity":"journal-of-infrared-millimeter-and-terahertz-waves","isNatureJournal":false,"hasQc":true,"allowDirectSubmit":false,"externalIdentity":"ijim","sideBox":"Learn more about [Journal of Infrared, Millimeter, and Terahertz Waves](http://link.springer.com/journal/10762)","snPcode":"10762","submissionUrl":"https://submission.nature.com/new-submission/10762/3","title":"Journal of Infrared, Millimeter, and Terahertz Waves","twitterHandle":"","acdcEnabled":true,"dfaEnabled":true,"editorialSystem":"em","reportingPortfolio":"Springer Hybrid","inReviewEnabled":true,"inReviewRevisionsEnabled":false}}],"origin":"","ownerIdentity":"b9a1db9b-c0d0-4312-97f7-75e5cec71105","owner":[],"postedDate":"December 19th, 2022","published":true,"recentEditorialEvents":[],"rejectedJournal":[],"revision":"","amendment":"","status":"published-in-journal","subjectAreas":[],"tags":[],"updatedAt":"2023-10-16T20:41:31+00:00","versionOfRecord":{"articleIdentity":"rs-2373644","link":"https://doi.org/10.1007/s10762-023-00917-2","journal":{"identity":"journal-of-infrared-millimeter-and-terahertz-waves","isVorOnly":false,"title":"Journal of Infrared, Millimeter, and Terahertz Waves"},"publishedOn":"2023-04-13 20:28:51","publishedOnDateReadable":"April 13th, 2023"},"versionCreatedAt":"2022-12-19 15:34:14","video":"","vorDoi":"10.1007/s10762-023-00917-2","vorDoiUrl":"https://doi.org/10.1007/s10762-023-00917-2","workflowStages":[]},"version":"v1","identity":"rs-2373644","journalConfig":"researchsquare"},"__N_SSP":true},"page":"/article/[identity]/[[...version]]","query":{"redirect":"/article/rs-2373644","identity":"rs-2373644","version":["v1"]},"buildId":"WrCJVZZCHTDjtuVLN7oU0","isFallback":false,"isExperimentalCompile":false,"dynamicIds":[84888],"gssp":true,"scriptLoader":[]}